Amplifying circuit

ABSTRACT

In signal sources having a high impedance, typically a capacitive “signal source” such as capacitor-microphone capsules, it is common practice to use amplifier circuits that include means for coupling signals and determining operating points in addition to the actual amplifier having a high-resistance, non-inverting input. For setting the operating points of the signal source and the amplifier, separate bias-voltage sources are provided; these are coupled to the signal source and the non-inverting input, respectively, of the amplifier via a coupling impedance. At least one coupling capacitance is disposed in the signal path between the signal source and the non-inverting input of the amplifier. To attain a considerable noise gain without the disadvantage of very high idle times in this type of amplifier circuit, it is proposed that the coupling impedances be formed from a nonlinear resistance (D 1 , D 2  or D 3 , D 4 ) and an ohmic resistance (R 3  or R 4 ) connected thereto in series. Moreover, the output signal (S 2 ) of the amplifier (ICI) or a signal derived therefrom is fed back to the coupling impedances (D 1 , D 2 , R 3  or D 3 , D 4 , R 4 ) via a second or third coupling capacitance (C 3  or C 4 ).

[0001] The invention relates to an amplifier circuit as defined in thepreamble to claim 1. An amplifier circuit of this type is known fromU.S. Pat. No. 3,595,998 A.

[0002] Usually, high-resistive resistances are used for setting theoperating point at high-resistance amplifier inputs, and forbias-voltage coupling to capacitive signal sources. FIG. 1 illustrates acorresponding circuit arrangement in accordance with the prior art.Here, C1 represents a capacitive signal source of 50 pF in the form of acapacitor-microphone capsule, which is coupled to a supply DC Vbias1 of+60 V via a high-resistive resistance R1 of 3 GOhms. The useful signalS1 of the capacitive signal source C1, for example an audio signal, issupplied to the high-resistance, non-inverting input (+) of an amplifierICI via a series-coupling capacitance C2 of 1 nF (which is inserted intothe signal path for separating the operating-point voltages). The outputsignal S2 of the amplifier ICI is fed back to the inverting input (−) ofthe amplifier ICI in the manner of negative feedback. In terms of thesignal voltage, this results in an amplification of V=1, making

[0003] available the output signal S2 with a low source impedance, whichcarries the same useful-signal information as the signal S1 with respectto value and phase. A bias source Vbias2 of +5 V, which is coupled tothe non-inverting input (+) of the amplifier ICI via a high-resistanceresistance R2 of 3 GOhms, is provided for setting the operating point ofthe amplifier ICI.

[0004] A circuit arrangement similar to the one in FIG. 1 is known fromU.S. Pat. No. 3,595,998 A. This known amplifier circuit likewise has acapacitor-microphone capsule M as the capacitive voltage source; itsoperating point is determined by an ohmic resistance R_(V), which isconnected to a first bias-voltage source U_(P). To effect a decouplingbetween the first bias-voltage source Up and a second bias-voltagesource U_(B) in order to set the operating point of the downstreamamplifier, a coupling capacitance C_(K) is disposed in the signal pathbetween the capacitor-microphone capsule M and the gate electrode 3 ofthe amplifier FET. The operating point of the amplifier transistor isdetermined by a resistance-divider network comprising the ohmicresistances R_(j), R₂, R₃, R_(V) and a diode D. Power-supply voltages ofarbitrary polarity can be used for the bias-voltage source U_(B),because field-effect transistors generally have a symmetricalconstruction, so the source and drain electrodes exchange functionsdepending on the applied voltage—that is, the respective electrodehaving the more negative voltage (in an N-channel model) assumes therole of the source. The output signal is obtained symmetrically in thesame manner at the source and drain electrodes by the components R₄, R₅,C and Tr. Only the operating voltage at the gate electrode must beadapted as a function of the polarity of the power-supply voltage,because the operating voltage does not generally correspond to one-halfthe power-supply voltage. This is effected by the diode D in series withthe resistance R₂. In the event of a negative supply voltage, theoperating point for the amplifier FET is produced by the voltage dividerR1/R3. In this instance, the diode D is blocked and ineffective. In thecase of a positive power-supply voltage, an inverted-voltage-dividerratio is necessary. This is accomplished by making the diode Dconductive and connecting the resistance R2 in parallel to theresistance R3.

[0005] The minimal value of the coupling resistances R1, R2 or R_(V)that is theoretically necessary results from the desired lower limitfrequency of the useful signal S1 to be transmitted. For example, with alower limit frequency of 20 Hz and a signal-source capacitance of 50 pF,the resulting value of the coupling resistances R1, R2 or R_(V), whichoperate in parallel with respect to the load of the signal source C1, orR_(V)(and whose parallel switching is effective as a load of the signalsource) would be 160 MOhms. This type of resistance value generates avery high noise voltage, which is, however, reduced to the ratio of theresistance value of the parallel circuit comprising the two couplingresistances R1, R2 or R_(V) to the value of the impedance of thecapacitive signal source, corresponding to the voltage division. It isalso the case that, when the resistance value of the parallel circuitcomprising the resistances R1, R2 or R_(V) is increased by a specificfactor, the noise voltage is further reduced by this factor; incontrast, the noise voltage generated in the parallel resistances R1, R2or R_(V) only increases by the root of the named factor, in accordancewith known laws of physics. With respect to calculations, this meansthat a noise gain of 3 dB is attained with each doubling of theresistance value.

[0006] Unfortunately, this increase in resistance is associated with aconsiderable drawback: The time that passes from the switching of theoperating voltages Vbias 1 and Vbias 2 (switch-on of device), or theswitching of the bias voltage Vbias1 to the capacitive signal source forloading the source capacitance and the necessary coupling capacitance,also increases linearly. It is common practice to use resistance valuesof 1 to 3 GOhms. In conventional microphones, the resulting load timesor idle times are in a range of 10 to 15 seconds; in microphones havinganalog-digital conversion, they can be more than 30 seconds because ofincreased operating-point requirements. Nevertheless, a further increasein the resistance value with the goal of a noise gain would bedesirable, because an extensive overlap by other noise sources does nottake place until about 10 to 20 GOhms. It is also to be anticipatedthat, in the case of a further increase in the resistance value for R1,R2 or R_(V), in practice the coupled operating-point voltages becomeincreasingly imprecise at the signal source or the non-inverting input(+) of the amplifier because of increasingly frequent, unavoidableleakage currents.

[0007] It is further known from EP 0 880 225 A2 to feed the outputsignal of the amplifier back to the connecting point of a seriesconnection comprising a high-resistive series resistance of twoantiparallel diodes, the connection being provided for setting theoperating point of the amplifier. In the cited reference, this feedbackis accomplished by the fact that virtually no differential voltageresults at the two ends of the antiparallel diodes (FIGS. 2 through 5),so the detrimental capacitance parallel to the diodes remainsineffective. In the circuit according to EP 0 880 225 A2, the operatingpoint of the signal source is not set by way of a separate bias-voltagesource, so no coupling capacitance is present in the signal path betweenthe signal source and the amplifier. In this known circuit, therefore,there is no issue of the shortest possible charging time because of theabsent coupling capacitance.

[0008] The same can be said for the amplifier circuit according to U.S.Pat. No. 5,589,799 A, in which there is also no biased microphonecapsule as a signal source, and thus also no coupling capacitance in thesignal path between the signal source and the amplifier.

[0009] It is the object of the invention to attain a considerable noisegain in an amplifier circuit of the type mentioned at the outset,without having to allow for the disadvantages of very high idle timesand an excessive influence of leakage currents. Advantageous embodimentsand modifications of the amplifier circuit according to claim 1 ensuefrom the dependent claims.

[0010] The invention is based on the consideration of replacing thecoupling resistances R1, R2 with a network comprising the seriesconnection of a nonlinear resistance and a high-resistive couplingresistance. The coupling resistance that determines the load times ofthe source capacitance and the coupling capacitance C2 can haverelatively small dimensions, because the nonlinear resistance isconductive with a low impedance during the load times, and after thecharging of the source capacitance and the coupling capacitance, theresistance automatically assumes a high resistance value that isnecessary for improving the noise performance.

[0011] The invention is described in detail in conjunction with FIGS. 2through 5. Shown are in:

[0012]FIG. 2 an electrical, basic circuit diagram of an amplifiercircuit according to the invention;

[0013]FIG. 3 a further electrical, basic circuit diagram of an amplifiercircuit according to the invention;

[0014]FIG. 4 a circuit diagram of a preferred embodiment of an amplifiercircuit according to the invention; and

[0015]FIG. 5 an electrical circuit diagram of a further preferredembodiment of an amplifier circuit according to the invention.

[0016] In the basic circuit diagram shown in FIG. 2, in comparison tothe prior art illustrated in FIG. 1, the coupling resistance R1 isreplaced by the series connection of a nonlinear resistance in the formof a diode D1 and a high-resistive resistance R3. With this measure, thebias voltage Vbias1 is supplied via the component D1, which has anonlinear current-voltage characteristic and is connected in series withthe resistance R3. After the capacitances C1 and C2 have been charged orrecharged, virtually no more current flows, so the diode D1 is in theblocked state because no voltage is present between the two terminalends of the diode D1. The noise voltage generated in the diode D1 istherefore very small, and is based on small, unavoidable leakagecurrents in the whole circuit. In contrast, during the charging of thecapacitances C1, C2, the diode D1 becomes conductive, and the loadingcurrents and times are determined by the ohmic resistance R3 connectedin series, the resistance having a resistance value of only 10 MOhms,which is lower than the coupling resistance R1 according to FIG. 1 by afactor of 300.

[0017] Without any further measures, the conductive state of the diodeD1 would also be caused by the applied AC-voltage useful signal S1because of the diode's current-voltage characteristic in the pass range.This would cause a detrimental, and nonlinear, load of the capacitivesignal source C1, because the source has a very high impedance. Thediode D1 would also at least partially transmit the noise voltagegenerated in the coupling resistance.

[0018] To completely prevent these undesired effects, a capacitance C3is provided, which feeds the impedance-converted useful signal S2 fromthe output of the amplifier ICI (whose signal amplification is set atV=1) back to the connecting point between the diode D1 and theresistance R3. The consequence of this feedback is that the usefulsignal S2 is identical in phase and value at both ends of the diode D1,so the diode D1 is permanently held in the no-voltage, or blocked,state, completely independently of the amplitude of the useful signalS2. This results in an extremely high-resistant and low-noise supply ofthe operating voltage Vbias1 to the signal source C1. In contrast, thediode D1 is conductive when the capacitances C1, C2 are recharged. Therecharging time is determined solely by the time constant R3/C2, whichis very small in comparison to the prior art illustrated in FIG. 1. Thedimensioning of the time constant R3/C2 depends on the desired lowerlimit frequency at which the described feedback of the useful signal S2is still adequately effective. A selected limit frequency of, forexample, one-tenth of the lowest useful frequency to be transmitted (inaudio signals, generally 20 Hz) fulfills this condition withoutlimitation. Moreover, the capacitor C3 functions such that the noisevoltage generated by the coupling resistance is short-circuited to theoutput of the amplifier ICI, and therefore no longer appearsdetrimental.

[0019] It may happen in practice that the bias voltage Vbias1 of thesignal source C1 varies during operation, for example for the purpose ofchanging the sensitivity or the directional characteristic in amicrophone capsule. In this connection, the described operationalmechanism is not only significant with respect to charging thecapacitances C1, C2 after the bias voltage Vbias1 has been switched on,but also for the accelerated discharge when a smaller, or even negative,bias voltage Vbias1 is to be set. In this case, as shown in FIG. 3, afurther diode D2 is in an antiparallel connection with the diode D1, sothe described charging processes can take place in bipolar fashion.

[0020] The inventive measures explained in FIGS. 2 and 3 can also beused for setting the operating point of the amplifier or the impedanceconverter ICI. For this purpose, as shown in FIG. 4, the couplingresistance R2 in accordance with FIG. 1 is replaced by a networkcomprising the components R4, C4, D3 and D4. This network, again,comprises a series connection of two antiparallel diodes D3, D4, and anohmic coupling resistance R4 associated with them in series, as well asa coupling capacitance C4 for feeding the useful signal S2 back to theconnecting point between the resistance R4 and the diodes D3, D4. Theoperational mechanism described in conjunction with FIG. 3 is thuseffected similarly to that of the amplifier ICI.

[0021] In the event that the useful-signal amplification of theamplifier ICI is to be greater than 1, the embodiment according to FIG.5 should be used; here, the output signal S2 fed back to the invertinginput (−) of the amplifier ICI in the sense of reverse feedback isdivided by a resistance divider comprising the ohmic resistances R5, R6.For example, with a division ratio of 10:1, the resulting signalamplification is V=10, because a signal S3 that is identical in valueand phase (compared to the useful signal S1 at the non-inverting input(+) of the amplifier ICI) is always established at the inverting input(−) of the amplifier ICI. In the exemplary embodiment according to FIG.5, for feeding the signal back to the base of the capacitances C3 andC4, it is not the output signal S2 that is used, but the signal S3,which is derived from the output signal, namely divided by the divisionratio R5/R6.

[0022] Of course, components having a nonlinear current-voltagecharacteristic can be used instead of the diodes D1 through D4, such asLEDs, Zener diodes, etc.

[0023] In practice, the amplifier circuit embodied in accordance withthe invention is used to attain a signal-noise ratio that is improved by2 to 10 dB, depending on the selected evaluation curve for measuring thenoise signal. At the same time, the undesired charging and rechargingtimes are reduced to less than one second.

1. An amplifier for audio-frequency signals, having acapacitor-microphone capsule as the capacitive voltage source, which iscoupled to a first bias-voltage source (Vbias1) via a first couplingimpedance (R3, D1; R3, D1, D2) for determining its operating point;having an amplifier (ICI), which is coupled by its non-inverting inputto a second bias-voltage source (Vbias2) via a second coupling impedance(R4, D3; R4, D3, D4) for determining its operating point; and having acoupling capacitance (C2), which is disposed in the signal path betweenthe capsule and the amplifier (ICI), characterized in that the first andsecond coupling impedances (R3, D1; R3, D1, D2; R4, D3; R4, D3, D4)respectively comprise a series connection of a nonlinear resistance (D1;D1, D2; D3; D3, D4) and a first or second ohmic resistance (R3; R4)having a relatively low resistance value, with the nonlinear resistancein the signal path between the capsule and the amplifier (ICI) beingconductive during the charging times of the capacitance (C1) of thecapacitor-microphone capsule and the coupling capacitance (C2), so thesecharging times are determined by the low-resistance first, or ohmicsecond, resistance (R3, R4), and with the nonlinear resistance (D1; D1,D2; D3; D3, D4) in each coupling impedance assuming a high resistancevalue after each charging of the capacitance (C1), thecapacitor-microphone capsule and the coupling capacitance (C2) in thesignal path between the capsule and the amplifier (ICI), so thenonlinear resistance only generates a very small noise voltage in eachcoupling impedance, and the output signal (S2) of the amplifier (ICI) ora signal derived therefrom is fed back to the first and second couplingimpedances.
 2. The amplifier circuit according to claim 1, characterizedin that a diode (D1 or D3) is provided as the first and second nonlinearresistance, respectively.
 3. The amplifier circuit according to claim 1or 2, characterized in that an antiparallel connection of two diodes(D1, D2 or D3, D4) is provided as the first and second nonlinearresistances.
 4. The amplifier circuit according to one of claims 1through 3, characterized in that the output signal (S2) of the amplifier(ICI) is fed back to the inverting input (−) of the amplifier (ICI), andto the first and second coupling impedances (D1, D2, R3; D3, D4, R4) viaa second and a third coupling capacitance (C3 or C4), with the fullvalue of its amplitude (FIG. 4).
 5. The amplifier circuit according toone of claims 1 through 3, characterized in that the output signal (S2)of the amplifier (ICI) is fed back to the inverting input (−) of theamplifier (ICI), and to the first and second coupling impedances (D1,D2, R3; D3, D4, R4) via a second and a third coupling capacitance (C3 orC4), with a partial value (R5/R6) of its amplitude (FIG. 5).